OPA603 Simplified Schematic

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Samuel Groner

Well-known member
Joined
Aug 19, 2004
Messages
2,940
Location
Zürich, Switzerland
Hi

I investigated the OPA603 simplified schematic (opa603.pdf) and came to the conclusion that there must be something wrong with it; the complementary common-emitter amplifier (the two transistors that have the collector going to one of the two diodes) seems to have ill-defined and very high bias currents.

Any hints on how to make a workable discrete implementation out of this are very much appreciated!

Samuel
 
> OPA603 simplified schematic ...there must be something wrong with it

Yeah. It is over-simplified.

> ill-defined and very high bias currents.

Yup.

> how to make a workable discrete implementation

First: forget discrete; this type of plan works much better if all devices are baked together so they match, and if they are all at the same temperature.

Even then, it will runaway when hot. The simplest un-simplification is to give the input current sources a very strong negative tempco, so the drop in the resistors matches the Vbe of the CE stages tied to the rails. But that gives each stage and most of the chip an idle current that drops a lot when hot, and that can't be a good idea.

Most discrete implementations of this general topology will have significant resistors. Put a resistor in the CE stage emitter that drops 1V: now you can bias it up pretty stable. But gain may be reduced and output swing will be less.

Another idea (and they may really do this in the 604) is to replace the resistor and CE with a current mirror, possibly one with gain more than 1:1. Now the currents are mirrored to the input stage current sources, very stable.

There is a large class of AB/B loudspeaker output stages that look like the second, third, and maybe fourth stages here, except the second stage emitters are split apart and get separate feedback from separate resistors in top and bottom of the output stage. The unity-gain version is very common. Dan Meyer did it with gain of 2 or 3, though this is tricky (some of his SWTP Tigers blew up all the time, others have run 30 years).
 
Thanks, PRR. Do you have a schemo of a Dan Meyer amp at hand?

I should add that I'm looking for a high-quality unity-gain discrete buffer for line-driving apps, squeezable into a 2520 footprint and flexible with supply voltage.

I already played with both your suggestions (emitter resistor and current mirror) before posting here but was not able to get an amp which simulates well - must be too stupid. I'll do some more research before I ask again...

However, I found a promising topology (SmithCFBamp.pdf, fig. 1). Currently things look like this (R4 is selected for lowest offset): [removed]

Hope you let me ask a few Qs on this topology (all refering to my schemo):
* what is the input impedance? 10k or very high? I'd vote for the latter, but not sure...
* how do I best compensate this amp? C1/C2 and R1 were found by trial and error and it looks to me as if stability and o/l-gain is still beta-dependent. I found it funny that R1 does and the impedance of the feedback network does not have any first-order influence on stability.
* current limiting is rather ugly and early - any simple fix for this? Probably not without an additional driver stage. R9/R10 are high-value for reasonable dissipation with +/- 40 V supplies.
* how about linearity? IISR (if I simulated right) o/l-gain is about 80 dB over the audio band which should make rather low THD.
* in the mentioned paper they call this a "switching amp"; what is switching here? Of course the output gets to AB somewhere, but this seems obvious. The other stages don't switch, do they?

Thanks very much!
Samuel
 
> what is the input impedance? 10k or very high? I'd vote for the latter, but not sure...

Like recent Presidential elections (I'm not just looking at Bush), electrons don't pay attention to the vote.

The dumb way to find out is to put a current probe on V3. If you can plot equations, plot something like V(V3)/I(V3) for an AC sweep. My vote is for very low I(V3) at DC and audio: where can current come into the right end of the 10K resistor? Q1 Q2 don't make any AC current. Q3 and Q4 have base currents a function of signal, but if Q3 Q5 Q7 and the other side have the large current gain you expect from a triple cascade, it should approach zero. I say 10Meg input impedance, in parallel with Q1-Q4 collector impedances, but sweep up to 100MHz: you will (probably) see some wild bumps.

I don't know why it current-limits: it looks like it should be a ~10 ohm resistor to the rail. There is a mechanism that will choke at ~1A, but that's an insane power for +/-40V and 600 ohm load.

> funny that R1 does and the impedance of the feedback network does not have any first-order influence

I don't see the feedback net, unless it has degenerated to a wire.

The Q3 Q4 stage's high frequency gain is mostly just C1/R1, and will be (one of) your dominant pole(s).
 
Samuel I would look at what your application's source impedance(s) is/(are) and make the amp's compensation take that into account. You are incurring a significant noise penalty by throwing 10k at the input, although in many applications you're probably dominated by noise from the source. The 10k does have the advantage of making things a bit more bulletproof at least.
 
[quote author="Samuel Groner"]I should add that I'm looking for a high-quality unity-gain discrete buffer for line-driving apps, squeezable into a 2520 footprint and flexible with supply voltage.
Samuel[/quote]

This sounds really worthwhile doing please don't give it up.
As yourself I like the idea of separating voltage gain stages from the final current gain stage that may drive a transformer. It seems silly to keep using the same hefty opamp inside circuits where they see 2k to 10k+ impedance on their outputs most of the time.
I was experimenting with a FET based Borbely-like opamp at unity gain. I tried replacing the output with BJTs and MOSFETs, but that was not a buffer, anyhow. Unfortunately, I have nothing useful to add besides a little encouragment.

Cheers,
Tamas
 
[quote author="Samuel Groner"]... I'm looking for a high-quality unity-gain discrete buffer for line-driving apps, squeezable into a 2520 footprint and flexible with supply voltage.[/quote]

Have you considered a CFP design?

Such as described here: http://www.dself.dsl.pipex.com/ampins/discrete/cfp.htm
 
Thanks for the answers.

I don't see the feedback net, unless it has degenerated to a wire.
Yep, straight wire (without gain :razz: ). A voltage divider sets this amp to more than unity gain, but DC offset gets ugly.

I don't know why it current-limits.
I don't see the mechanism involved either, but in simulation it's somewhere at 250 to 500 mA.

You are incurring a significant noise penalty by throwing 10k at the input.
Making R1 smaller and C1/C2 larger seems not to work well, as I get some slew-limiting (rather fast, though). As shown, large-signal BW looks larger than small-signal.

This sounds really worthwhile doing please don't give it up.
I'm not giving up, just too much other things to finish! In fact, I believe that the design is ready for breadboard as shown - if someone is faster than me, feel free to go ahead!

Have you considered a CFP design?
Yes, but I wasn't sure how to convert this to a push-pull stage. I don't like SE stuff for outside-world-interfacing.

Would the Jung buffer be of any use?
I don't think so - the diamond buffer need's to be used inside of a feedback loop for decent performance. This one should be stand-alone.

Samuel
 
A few notes to add:
* Q1-Q6 can be about any good complementary small signal transistors; 2N4401/2N4403, BC550/BC560, 2SA1015/2SC1815 or 2N3904/2N3906 as a few suggestions. Supply voltage may need to be lower as shown.
* Q7/Q8 are MJE171/MJE181 (or one of the other voltage ratings as appropriate). As a replacement consider the BD139/BD140 family.
* C1/C2 are ceramic C0G/NP0
* replace R4 with a 680 ohm and parallel it with one for lowest output offset selected resistor. Selection should be done with intended supply voltage.
* for improved PSRR and stability, add a RC network (10r0 and 1 uF stacked film to ground) between R9 and Q7 collector as well as R10 and Q8 collector.
* use 220 uF in parallel with 100 nF for power supply decoupling.
* to isolate capacitive loads, use a JT-OLI-3 or a small resistor (39r0 minimum) between output and load. The former is prefered when driving an output transformer.

Samuel
 
[quote author="Samuel Groner"]

Have you considered a CFP design?

Yes, but I wasn't sure how to convert this to a push-pull stage. I don't like SE stuff for outside-world-interfacing.

[/quote]

See figure 13 on the page I indicated. That is a push-pull design!
 
Right, what I wanted to say is that I'm looking for a topology that switches to class AB under heavy loads. This is not the case here, is it? Never studied these non-complementary designs in detail, so not sure about it.

Or is "push-pull" equivalent to "switching to AB"?

Samuel
 
I don't think "push-pull" says anything useful regarding class. It has been used most of the times to say complementary topology, but it does not convey any idea about operating mode of the output stage.

I took a second look at the schematics and now I grok a little better what it does. At least I think I do. Correct me if I am wrong.
Q3, Q5, Q7 form a CFB amplifier that is run at unity gain in non-inverting mode. Q4, Q6 and Q8 are the complementary of the top circuit.
If so the upside is that it has good bit of gain inside that loop for error correction. This may help clean things up a bit when the load draws more than the standing current can give.
D. Self says that CFP is the way to go when driving low impedances. Then again following him to the word will make you end up with a two hundred transistor amplifier. Also, CFPs can be trickier than followers as they have gain and can have all the ills that come with gain like instability.

Going back to the problem on hand. (don't ever use the phrase "defining the problem domain" on this board, someone tried it before and was fed to the lions) Is the worst case scenario to drive a 75 ohm primary?
How many dBs do you want to be able to drive into the lowest impedance you want? Do you have power supply limitations, rail voltages, current? Are there any space limitations or restrictions on heat output?
Peraps answering those questions will help figure out the approach you take or keep.
 
[quote author="tk@halmi"]Self says that CFP is the way to go when driving low impedances. Then again following him to the word will make you end up with a two hundred transistor amplifier. Also, CFPs can be trickier than followers as they have gain and can have all the ills that come with gain like instability.[/quote]

I'm not taking all that Self writes as the gospel word of amplifier design but he does back up his claim with facts and his reasoning seems both sound and logical.

I don't know about any 200 transistor amps but this push-pull CFP seems very well thought out to me.

Gain = instability? Oh well... :roll:
 
[quote author="cuelist"]I'm not taking all that Self writes as the gospel word of amplifier design but he does back up his claim with facts and his reasoning seems both sound and logical.[/quote]
Sometimes he is. You should make this buffer with a CFP output then.
 
Tamas, this thread/design started a bit off-topic, that's why I didn't stated my design goals clearly. But of course you're right so I'll add them now:
* easy to use with different power supplies between +/- 15 V and +/- 24 V, preferably up to +/- 40 V
* offset < 20 mV any time (for easy use with servos)
* class A drive into 600 ohm @ 40 Vpp with excellent linearity
* current limited to about 250 mA
* reasonable freedom of slewing
* excellent stability when driving difficult loads
* input impedance 10k or higher
* reasonable PSRR
* easy to build (i.e. no semiconductor matching and selecting)
* easy to source parts only
* 2520 footprint

What I suspect without building it is that the design as shown does meet all the above points.

What I don't like about it is that I don't fully understand it. Unfortunately, I can't find the original dissertation here in Switzerland to learn more about it.

You're right that it is a complementary CF amp. However, I believe(d) that all CF amps need a feedback resistor for stability - which is not the case here. In turn the source impedance (usually mainly R1) is part of the compensation. The R4/R5 node should be low impedance, so why doesn't it load the output?

I thought about making Q7/Q8 CFPs for better linearity, but open-loop gain is about 80 dB over the full audio band, so linearity should be pretty good already.

Samuel
 
[quote author="Samuel Groner"]

The R4/R5 node should be low impedance, so why doesn't it load the output?


Samuel[/quote]

It does, but calculate by how much. The load is dependent on the source Z in series with the 10k, times beta. If these were reduced eventually you get to the r sub e's and R5/6, which are effectively in parallel.

But with feedback things equilibrate quickly in any case. The required current to keep the loop closed at d.c. to moderate frequencies is small, given the fairly high Z output node, which is dominated by the Q7/8 beta times the load resistance---maybe 60k-120k depending, with the 600 ohm load. With light loading eventually the impedances at the collectors of Q5 and Q6 will start to limit the max Z.
 
> Or is "push-pull" equivalent to "switching to AB"?

No, but in audio it goes-together-well.

You can't pass audio through Class C.

You can't build a single-ended Class B audio amp.

Class C and SE Class B are only good for crude power or with tuned loads.

So single-ended audio amps MUST be Class A.

Push-pull audio amps can also run in Class A. But in push-pull, Class B audio becomes possible, and the advantages of Class B are compelling. Power loss is half of Class A at full power: 3/4 the power iron, 1/2 the heatsink. In speech/music operation, the advantage is greater, since we run below 10% max power 90% of the time. A Class B speech/music amp will run cool 90% of the time, a Class A amp idles at twice the max sine power 90% of the time. And believe it or not, Class AB can give lower THD than Class A (but it is easy to lose this advantage, and THD is a terrible audio metric).

Text-book Class B requires the devices to be cut-off at zero signal. That's unrealistic. In speech/music systems, we never have zero signal, we have a dynamic range. There is always something down at the bottom of the range, and it may be important. Also all practical devices lose gain at low current, so for low-distortion operation we have to bias them up to some minimum current that gives the same/similar gain for small or large signals. In "small" (up to 6L6) tubes, "Class AB" may be more A than B, reaching cut-off only on the largest peaks. In BJT amps, good linearity may be had with idle currents less than 1/10th of peak current. In almost all normal operation, the devices go to cut-off (or a standby bias) pretty much all the time (each half-cycle except the softest passages).

---
> I didn't state my design goals ... I'll add them now:

At what gain??

You show unity-gain, but then you mention "2520 footprint". IIRC, the 2520 has a HIGH impedance inverting input, for flexible gain setting and relative indifference to feedback network impedance. This plan seems to have a low impedance inverting input (assuming you break the unity-gain feedback link). It will "work" in voltage-feedback schemes, but performance will vary with feedback network impedance, which is less true for voltage feedback amps like 2520.

The obvious way to meet your goals, stated and implied, is the good old 990 topology, also seen in Self and probably in the 2520. Yes, it isn't push-pull throughout. As you found, setting bias current is a lot easier if one side varies and the other side is a fixed current source, because overall feedback can force the varying side to balance the fixed current. The diff-pair input stage naturally tends to high impedance both sides and to low offset voltage (indeed if it works at all, it probably works with <20mV offset). You can adopt/adapt the 990 values and meet the 40Vpp 600Ω and 250mA specs; you can simplify the 990's complex compensation if you only work at modest gain and/or don't wish to plagiarize too much.

The Diamond Buffer can work outside a feedback loop and can push somewhat Class AB with excellent linearity. The main flaw against your stated goals is that DC offset is sure to be >=+20mV (because PNP stuff don't conduct as well as NPN stuff), and there is no good place to trim/servo that out.

40Vpp 600Ω on one hand, 250mA on the other hand, begs the question: what does it do between 33mA and 250mA? Do you simply want to keep potential destruction down to 1/4 Amp, distortion not specified, or do you want to drive <100Ω loads cleanly? (And if so, at 40Vpp/2 Watts, or will you take less?)
 
At what gain?
Noninverting unity gain.

2520 footprint
Just a convenient industry standard - inverting input would be NC.

The obvious way to meet your goals, stated and implied, is the good old 990 topology, also seen in Self and probably in the 2520.
Well, this topology has it's drawbacks at unity gain, i.e. exceeding of CM input range and typically rather low slew rate. A folded cascode would perhaps be more suited for this task if we want a voltage feedback amp, but these designs tend to be more complicated.

What does it do between 33mA and 250mA?
250 mA is simply the limit where MJE171/MJE181 don't die without heatsinking when driving a short. No intention to drive anything below 600 ohm.

Samuel
 
BTW: Just because the basic circuit doesn't have a feedback R doesn't mean it might not be helpful for transient response.

In fact, it looks like something around 300 - 1k allows one to dispense with any other compensation and be much less source-Z sensitive. The bandwidth is affected by source Z since there is a capacitive component around 10pF, fine for most audio apps. Gain is about 0.997 for a 100 ohm source Z and driving 600 ohms; if you must have precisely unity you can add a trim R from the junction of the feedback R, R5 and R6, to common, to trim it up.
 

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